Method and apparatus for mitigating adverse effects of bonding wire of external optical modulators

ABSTRACT

An optical transceiver including a submount, a Mach-Zehnder Modulator (MZM), bonding wires, and a low pass filter type matching network is provided. The MZM includes an input port and an output port and disposed on the submount. The bonding wires are coupled to the submount and the MZM. The low pass filter type matching network is coupled to the bonding wires and is configured to absorb inductance of the bonding wires at a high frequency.

CROSS-REFERENCES AND RELATED APPLICATION(S)

This application is a divisional of U.S. patent application Ser. No.16/554,464 filed Aug. 28, 2019, which claims priority to U.S.Provisional Application No. 62/724,890, filed Aug. 30, 2018, titled“METHOD AND SYSTEM FOR MITIGATING ADVERSE EFFECTS OF BONDING WIRE OFEXTERNAL OPTICAL MODULATORS,” the contents of which are incorporatedherein by reference in their entirety.

TECHNICAL FIELD

The present disclosure relates to designs and techniques for mitigatingeffects of bonding wires, and in particular to the techniques formitigating the effects of bonding wires of external optical modulatorsincluding Mach-Zehnder Modulators on a submount.

BACKGROUND

Bonding wires comprise an internal part of integrated circuit (IC)packaging for making connections to other circuitry, such as activedevices including optical modulators and for input and outputconnections. As such, bonding wires are used extensively in packagingtechnology for chips. However, the bonding wires introduce extraparasitic inductance in the form of inductance in series with resistanceat high frequencies. Further, the number of bonding wires, their heightsfrom the substrate, frequency and dimension may often play an importantrole in overall circuit performance.

In the design of an optical communication system, for example, aMach-Zehnder Modulator (MZM) is often used in the optical communicationsystem. For data modulators, a semiconductor MZM is a preferredmodulator design because it may be integrated with a tunable laser, andbecause of optical data modulation characteristics, low electrical drivevoltage requirements, compact size and programmable transmissioncharacteristics. Integrated transmitters are fabricated by couplinglight from the primary laser output mirror. In the case of the MZM, themodulator input is split up into two optical waveguide paths ormodulator arms, and then combined into a common data modulated outputwaveguide and a secondary waveguide that can be used for opticalmonitoring. Data is modulated onto the tunable laser output by drivingone or both of the MZM arms with an electronic data signal that affectsthe physical properties of the MZM waveguides via electrical electrodesor interconnects.

When the MZM is mounted on a submount, bonding wires are used to connectinput and output ports of the MZM to the submount. The bonding wireshave a typical length of several hundred micrometers and ischaracterized by an inductance, e.g., inductance of about 0.2-0.3 nH andmay cause many disadvantageous effects such as signal degradation. Forexample, the inductance of the bonding wires causes radio frequency (RF)attenuation at high frequencies, specially when a bandwidth of the MZMexceeds 10 GHz. Another problem associated with the inductance of thebonding wires is that the impedance of the MZM is usually lower than 50Ohm, and thus seriously affect a high speed MZM packaging.

Therefore, there is a need for new and improved techniques formitigating or reducing the disadvantageous effects of the bonding wirescausing signal degradation in the design of an optical communicationsystem.

SUMMARY

According to the present disclosure, an optical transceiver is provided.By way of example, an optical transceiver includes a submount, aMach-Zehnder Modulator (MZM) including an input port and an output port,bonding wires coupled to the submount and the MZM, and a low pass filtertype matching network coupled to the bonding wires. The low pass filtertype matching network is configured to absorb inductance of the bondingwires at a high frequency.

In an aspect of the present technology, the low pass filter typematching network may include a 3^(rd) order Butterworth filter.

In another aspect of the present technology, the low pass filter typematching network may include a first matching network coupled to theoutput port of the MZM and the first matching network may include aresistor, an inductor, and a capacitor and be configured to absorb theinductance of the bonding wires.

In another aspect of the present technology, the low pass filter typematching network may include a second matching network coupled to theinput port of the MZM and the second matching network may include aninductor, and a capacitor and be configured to absorb the inductance ofthe bonding wires. Further, the second matching network may furtherinclude a resistance in the form of input impedance.

In another aspect of the present technology, the low pass filter typematching network may include the first matching network coupled to theoutput port of the MZM and the second matching network coupled to theinput port of the MZM.

In another aspect of the present technology, a method of implementing alow pass filter (LPF) type matching network configured to absorbinductance of bonding wires at a high frequency between an externaloptical modulator and a submount in a packaging of the external opticalmodulator is disclosed.

In another aspect of the present disclosure, the method of implementingthe LPF type matching network may comprise providing a first matchingnetwork coupled to an input of the external optical modulator and asecond matching network coupled to an output of the external opticalmodulator.

Further, in another aspect of the present disclosure, in the method, theexternal optical modulator may comprise a Mach Zehnder Modulator (MZM).Further, the high frequency may comprise a frequency greater than 10 GHzand the LPF type matching network may comprise a 3^(rd) orderButterworth filter.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features, aspects and advantages of the presentdisclosure will become better understood from the following description,appended claims, and accompanying figures where:

FIG. 1 is a diagram conceptually illustrating an embodiment of thepresent technology in accordance with an aspect of the presentdisclosure;

FIG. 2 is a diagram conceptually illustrating an embodiment of thepresent technology in accordance with an aspect of the presentdisclosure;

FIG. 3 is a flowchart conceptually illustrating an embodiment of thepresent technology in accordance with an aspect of the presentdisclosure;

FIGS. 4A and 4B are examples of a frequency response and correspondingSmith Chart conceptually illustrating various aspects of an embodimentof the present technology in accordance with an aspect of the presentdisclosure;

FIGS. 5A and 5B are examples of circuit diagrams conceptuallyillustrating an embodiment of the present technology in accordance withan aspect of the present disclosure;

FIG. 6 is an example of a frequency response conceptually illustratingan embodiment of the present technology in accordance with an aspect ofthe present disclosure;

FIG. 7 is an example of a circuit diagram conceptually illustrating anembodiment of the present technology in accordance with an aspect of thepresent disclosure;

FIG. 8 is an example of a circuit diagram conceptually illustrating anembodiment of the present technology in accordance with an aspect of thepresent disclosure;

FIG. 9 is an example of a circuit diagram conceptually illustrating anembodiment of the present technology in accordance with an aspect of thepresent disclosure;

FIGS. 10A-10F are examples of simulation results including frequencyresponses, a Smith Chart, and a group delay response conceptuallyillustrating various aspects of the present technology;

FIG. 11 is an example of a circuit diagram conceptually illustrating anembodiment of the present technology in accordance with an aspect of thepresent disclosure;

FIG. 12 is an example of a circuit diagram conceptually illustrating anembodiment of the present technology in accordance with an aspect of thepresent disclosure;

FIGS. 13A-13D are examples of simulation results including frequencyresponses, a Smith Chart, and a group delay response, conceptuallyillustrating various aspects of the present technology in accordancewith an aspect of the present disclosure;

FIG. 14 is an example of a circuit diagram conceptually illustrating anembodiment of the present technology in accordance with an aspect of thepresent disclosure; and

FIGS. 15A-15D are examples of simulation results including frequencyresponses, a Smith Chart, and a group delay response conceptuallyillustrating various aspects of an embodiment of the present technologyin accordance with an aspect of the present disclosure.

DETAILED DESCRIPTION

The detailed description of illustrative examples will now be set forthbelow in connection with the various drawings. The description below isintended to be exemplary and in no way limit the scope of the presenttechnology. It provides a detailed example of possible implementationand is not intended to represent the only configuration in which theconcepts described herein may be practiced. As such, the detaileddescription includes specific details for the purpose of providing athorough understanding of various concepts, and it is noted that theseconcepts may be practiced without these specific details. In someinstances, well known structures and components are shown in blockdiagram form in order to avoid obscuring such concepts. It is noted thatlike reference numerals are used in the drawings to denote like elementsand features.

Further, methods and devices that implement example embodiments ofvarious features of the present technology are described herein.Reference in the description herein to “one embodiment” or “anembodiment” is intended to indicate that a particular feature,structure, or characteristic described in connection with the exampleembodiments is included in at least an embodiment of the presenttechnology or disclosure. The phrases “in one embodiment” or “anembodiment” in various places in the description herein are notnecessarily all referring to the same embodiment.

In the following description, specific details are given to provide athorough understanding of the example embodiments. However, it will beunderstood by one of ordinary skill in the art that the exampleembodiments may be practiced without these specific details. Well-knowncircuits, structures and techniques may not be shown in detail in ordernot to obscure the example embodiments (e.g., circuits in blockdiagrams, etc.).

In packaging, bonding wires comprise an internal part of integratedcircuit (IC) packaging for making connections to other circuitry such asactive devices including optical modulators and for input and outputconnections. As such, bonding wires are extensively used in packagingtechnology for chips. However, it is well known that the bonding wiresintroduce extra parasitic inductance in the form of inductance in serieswith resistance at high frequencies. Further, the number of bondingwires, their heights from substrate, frequency and dimension may oftenplay an important role in overall circuit performance. As such, extrainductance introduced by the bonding wires is considered as the mainparameter of the bonding wires and at higher frequencies the extrainductance becomes a factor affecting the overall system performance,e.g., often degrading system performance.

The present technology disclosed herein provides solutions addressingthe problems noted relating to the bonding wires in high speedpackaging. In one aspect of the present disclosure, for illustrationpurposes of the present technology, in one example, a Mach-ZehnderModulator (MZM) 101 including input ports and an output port may bedisposed on a submount 105 as shown in FIG. 1 . The MZM 101 may betypically driven by a differential driver and may compose a set of twooptical waveguides 103 running in a semiconductor die chip and a set oftwo electrodes running on top of the two waveguides. In the example, oneoptical waveguide may be split into two optical waveguides 103 which areshown in the example in FIG. 1 . The optical signals running in the twooptical waveguides 103 may be then phase modulated in each arm of themodulator section 101 by differential electrical signals applied at eachinput port 102 of the MZM 101. The two phase-modulated signals may thenbe combined into one output optical waveguide. The two differentiallyphase modulated optical signals are combined in a way of in-phase orout-of-phase. In this way, the combined optical signal may result in anamplitude modulated signal. Also, the bottom of the MZM chip may bemetallized and attached to a ground plane of the submount 105.

Generally, for high speed applications, the performance of a device maybe limited by the electrical parasitic capacitance, series resistance,and bonding wire inductances. Especially, in the example, the inductanceof the bonding wires 107 at the input port 102 of the MZM 101 adverselyaffect a frequency response of the MZM's gain, degrading the overallperformance of the device. Therefore, the elimination or reduction ofthe adverse effect of the bonding wires such as the bonding wires 107 inthe example is very desirable ad is important in improving theperformance of a MZM response of the device.

As mentioned above, the MZM 101 shown in FIG. 1 may be implemented on asubstrate and thus on a chip. The MZM 101 may be coupled to one or moretransmission lines 109 at which an input signal may be applied. Also,one or more bonding wires 107 may be used to couple the one or moretransmission lines and the MZM 101 via input ports 102. Further, anoutput port may be terminated with a passive network 111 includingelements representing resistance (R), inductance (L), and/or capacitance(C). In the example of FIG. 1 , a dotted circle indicates the passivenetwork 111 coupled to the output port of the MZM 101.

FIG. 2 provides a more detailed view of one side of the MZM of FIG. 1 ,in an aspect of the present technology, conceptually illustrating oneimplementation of inductance and/or capacitance on the chip. That is,the inductance and/or capacitance on the chip may be implemented oradjusted by changing a shape of the electrodes. Also, by adjusting thelength of the bonding wire 107, the inductance may be modified. Further,by combining on-chip inductance, capacitance and bonding wireinductance, various types of matching network, for example the matchingnetwork 111, may be implemented at the input port and/or output port ofthe MZM 101. Further, the termination resistances may be implementedeither by discrete resistors or on-board resistors embedded on thesubmount 105. Furthermore, the size of a pad of the resistor may beacting as a capacitor for the matching network 111 and thus thecapacitance may be varied by adjusting the size of the pads.

In an aspect of the present disclosure, by utilizing the characteristicsof a low pass filter (LPF), an input and/or output matching network 111for MZM 101 may be designed to eliminate or reduce the adverse effectsof parasitic elements including the inductance of bonding wires 107.Thus, modulation signals that is to be terminated may be fed back to theMZM 101, thereby improving the frequency response of the MZM 101 and asa result, the bandwidth of the MZM 101 may be significantly improved,and in one aspect of the present disclosure the effects of group delay(GD) caused by electrical components (e.g., bonding wires, etc.) may bemitigated significantly.

Referring back to FIG. 1 , the MZM may include a transmission linelocated or disposed above the optical waveguide 103. The opticalwaveguide 103 may include a p-n junction which is acting as a capacitiveloading to slow down a traveling speed of an electrical signal. In theexample, optical signals may be phase modulated as an electricalmodulating signal travels along the transmission line. In this way, thepropagation speed of the electrical signal may be matched to some extentwith the propagation speed of the optical signal. Further, in theexample, due to the capacitive loading effect of the p-n junction, thecharacteristic impedance of the transmission line is usually less than50 Ohms. Also, since the driving signal source is about 50 Ohm based, animpedance transformation (or impedance matching) may be implementedbetween the input 50 Ohm source impedance and the input impedance of thetransmission line of a modulator.

There are various approaches for implementing a wideband impedancetransformation. By way of example, one way to implement the widebandimpedance transformation is to use a taper design technique which givesa smooth impedance transition from 50 Ohm to the input impedance of amodulator. However, one expected drawback of this approach, for example,using the taper design technique, is that the impedance of a bondingwire between a taper implemented on a submount and a modulator chip maynot be included in the design of the taper. As a result, the impedanceof the bonding wire may significantly affect a frequency response of asystem when the designed modulator has a very wide bandwidth oroperating at a high frequency.

Further, in many different applications a wideband modulator is commonlyused with bandwidth of the wideband modulator exceeding far more than 20GHz, but with degraded performance due to inductance or impedance ofbonding wires. Therefore, in an aspect of the present disclosure, thepresent technology disclosed herein provides a technique or methodologyproviding a capability of absorbing the effects of the bonding wires.

In one implementation, in an aspect of the present disclosure, a lowpass filter concept may be utilized to reduce or mitigate the effect ofthe impedance of the bonding wire. That is, in an aspect of the presentdisclosure, a low pass filter (LPF) or the like may be implementedbetween the source impedance of 50 Ohm and the input impedance of amodulator in such a way that the LPF is configured to absorb theimpedance of the bonding wire as part of the LPF. That is, the LPF maybe designed to have a 50 Ohm source impedance and the input impedance ofthe modulator. Further, in an aspect of the present disclosure, thebandwidth of the LPF is much wider than the bandwidth of a modulatorchip. Further, to improve the system performance, the termination of themodulator needs to be dealt with. By controlling the impedance of thetermination of the modulator, a frequency response, input impedance andgroup delay characteristics may then be controlled.

FIG. 3 shows an example flowchart conceptually illustrating an exampledesign methodology in an aspect of the present disclosure. By way ofexample, a design methodology for the termination of a modulator isprovided. In the example, an inductance-capacitance-inductance (L-C-L)low pass filter (LPF) circuit topology may be used to approximate amodulator or a modulator section. The transmission line on an opticalwaveguide may be any transmission line type such as a CPS, CPW ormicrostrip line. In an aspect of the present disclosure, thesetransmission line types may be approximated as a simple microstrip lineand may be approximated as a 3^(rd) order L-C-L lumped circuit.

In an aspect of the present disclosure, the above design methodology maybe applied to a Mach Zehnder Modulator design. As mentioned above, FIG.3 illustrates an example design flow in connection with the Mach ZehnderModulator design. At S301, first, a Mach Zehnder Modulator (MZM) ismodeled using a L-C-L low pass filter type (LPF) equivalent circuit. AtS303, a low pass filter is designed and selected to absorb an equivalentcircuit of a transmission line as part of the L-C-L LPF. At S305, a lowpass filter type matching network is synthesized to have a maximum −3 dBcut off frequency. At S307, the synthesized low pass filter network isimplemented to reduce or mitigate the effects of bonding wires of theMZM.

By way of example, in one implementation and in an aspect of the presentdisclosure, a 3^(rd) order low pass filter may be adopted to absorb anequivalent circuit of the transmission line as part of LPF. Further,since the modulator has a low impedance including high capacitance dueto a capacitive loading, a 3^(rd) order L-C-L LPF circuit topologyhaving a high capacitance value may be selected. In an aspect of thepresent disclosure, a Butterworth filter may be used for a candidateL-C-L LPF circuit topology. Butterworth filters have relatively highcapacitance compared to other types of filters. Also, in another aspectof the present disclosure, other types of filters (i.e., other thanButterworth filters) may be used as long as the equivalent circuit ofthe modulator can be absorbed in the LPF (i.e., the other types offilters). However, the designed bandwidth needs to be wide enough tomeet the required bandwidth of the modulator. Further, a designed LPF(i.e., a Butterworth filter) may be synthesized to have a maximum −3 dBcut off frequency, where the capacitance of the converted LPF is thesame as the one of the approximated 3^(rd) order equivalent circuit ofthe modulator. In this case, in an aspect of the present disclosure, theinductance of the synthesized LPF may be equal to or greater than thoseof the approximated equivalent circuit of the modulator. In anotheraspect of the present disclosure, the differences may be taken intoaccount when the termination and the input matching networks aresynthesized.

To further illustrate the present design methodology, designs detailsare provided below to give a better understanding of various aspects ofthe present technology.

Approximation of Modulator. From original S-parameters extracted from anMZM design, a modulator may be approximated as an L-C-L circuit. Forexample, input reflection curves of both original design s-parameter andextracted L-C-L circuit of the modulator are shown in FIG. 4A. In FIG.4A, original data (e.g., dBS11) is shown as a solid line and the inputreflection data (e.g., dBS99) of the extracted L-C-L circuit is shown asa dashed line. For illustration purposes, the MZM modulator that isdesigned is for 10 Gb/s, and thus the response curve of theapproximation (e.g., dBS99) relatively matches well with the originaldata (e.g., dBS11) up to about 10 GHz. Corresponding Smith Charts ofdBS11 (solid line) and dBS99 (dashed line) are shown in FIG. 4B.

Further, FIGS. 5A and 5B illustrates equivalent circuits approximatingthe MZM modulator. That is, FIG. 5A illustrates an equivalent circuitapproximating the MZM modulator and for the design of terminationimpedance using a low pass filter concept, FIG. 5B illustrates aschematic for the simplified circuit. FIG. 6 shows impedance of themodulator over a frequency range. In the example, it may be observedthat resistive values mainly affect the impedance at less than 2 GHzwhen the impedance of the modulator is observed. As such, it may benoted that the approximation of the MZM modulator is valid at above 2GHz.

Scaling to a low pass filter (LPF). FIG. 7 illustrates a synthesized3^(rd) order LPF in accordance with an aspect of the present disclosure.In an aspect of the present disclosure, it is noted that a Butterworthlow pass filter has relatively large normalized capacitance. As such, aButterworth low pass filter may be chosen to absorb an equivalentcircuit of a transmission line. By way of example, in case of a 3^(rd)order Butterworth LPF, the normalized values may be chosen such thatL1′=1, C2′=2, ad L3′=1 as shown in FIG. 5B. The normalized C2′ is thenscaled to a certain −3 dB frequency to have 0.575 pF with 50 Ohm sourceand load impedance, for example. Further, in an aspect of the presentdisclosure, an arbitrary source and load impedance may be chosen as wellto absorb the capacitance and the inductances. In the example shown inFIG. 7 , the −3 dB frequency of a Butterworth LPF obtained in this waymay be 11.077 GHz. The inductance values obtained for L1 and L3 may bethen 0.3981 nH. Thus, in the example, by adding, 0.3981 nH-0.275nH=0.123 nH, the Butterworth LPF having 11.077 GHz may be synthesized.

Modelling of MZM and Load Matching. FIG. 8 illustrates a synthesized lowpass filter (e.g., LPF1) between an output of the Butterworth filter(e.g., at an output of the MZM) as shown in FIG. 7 and a load R_(L) of50 Ohm. As shown in FIG. 8 , the low pass filter (e.g., LPF1) may besynthesized between the output of the Butterworth LPF and the loadR_(L).

In the example, by controlling a bandwidth (BW) of the LPF1, a responsecharacteristic of the modulator (MZM) may be modified. Further, in theexample, a reflected signal from the input of LPF1 (i.e., S11 of LPF1)is fed back to the modulator (MZM), and as a result, the fed back signalfrom the LPF1 modifies the S21 response and group delay of themodulator. This may be interpreted as follows. Some of the modulationsignal going to the load impedance (e.g., R_(L)) is fed back into themodulator with a slight delay and improves the modulation efficiency. Asa result, the frequency response of the modulator at high frequenciesmay be improved since the reflected signal at the high frequencies isgreater than signals at lower frequencies. It is further noted that ingeneral, the reflection from higher frequencies of LPF1 is greater thanthe one from lower frequencies.

Design of LPF for Termination. A low pass filter synthesis for LPF1between the Butterworth LPF and the termination (e.g., R_(L)) may becarried out. In an aspect of the present disclosure, the bandwidth (BW)of the matching network (e.g., LPF1) may be much wider than thebandwidth of the Butterworth LPF. It is because the reflected signal S11of the matching network LPF1 may be fed back into the Butterworth LPFand the reflected signal may contribute as a modulation signal. As aresult, it is observed that the S21 bandwidth of the Butterworth LPF iswidened and the group delay is flattened, that is, improved.

As such, it is important to choose an appropriate bandwidth of LPF1 andan appropriate filter type for the matching network LPF1. By way ofexample, in accordance with an aspect of the present disclosure, aBessel type filter having about 3 times BW of the Butterworth LPF (i.e.,30 GHz) may be selected, for LPF1. By selecting the bandwidth of LPF1 inthis way, some high frequency signal components may be fed back to theButterworth LPF and thus the high frequency response of S21 of themodulator (MZM) may be compensated.

In one example, FIG. 9 illustrates synthesized component values of themodulator, the Butterworth LPF, and the matching network LPF1. For thematching network LPF1, a Bessel type LPF may be chosen to have L4=0.0876nH, C2=0.103 PF, and L3=0.585 nH. Also, further optimization may becarried out by adjusting the bandwidth of the matching network LPF1. Inan aspect of the present disclosure, if the equivalent LPF circuit ofthe MZM has an arbitrary source and load impedance, the synthesis of thematching network LPF1 has to be done with the arbitrary output impedanceof the MZM LPF circuit. In the example, the inductance L2 may be part ofthe Butterworth LPF, and L2 is set to 0 for the purpose of simulatingthe modulator MZM response together with the termination on its rightside including L1, L4, L3, C1 and R1. Further, in the example, L2 may betaken into account later when the input matching network is synthesized.Also, port 4 (e.g., P4) impedance is set to 1000 Ohm to not affect thenetwork characteristics.

FIGS. 10A-F show various simulation results of the synthesized matchingnetwork LPF1 of FIG. 9 . In the simulation results, solid linesrepresent the responses of the original MZM data of the modulator MZMwith a 50 Ohm load, and dashed lines represent the responses with thesynthesized matching network LPF1 and a 50 Ohm load. As can be seen inFIG. 10A and FIG. 10F, the response of dBS11 is shown to have improvedup to 10 GHz and the group delay (GD) of the system has significantlyimproved with the synthesized matching network LPF1 (e.g., less than 7ps compared to about 13 ps).

FIG. 10A shows input reflection responses (dBS11, dBS33). In theexample, it may be noted that up to 12 GHz, the input reflection hasbeen improved but it has slightly degraded at above 12 GHz. FIG. 10Bshows a corresponding Smith Chart, further illustrating that the inputreflection has improved as well at lower frequencies. FIG. 10C showsthat the input impedance of the modulator MZM has increased slightly upto 13 GHz, which corresponds to FIGS. 10A and 10B, where the inputreflection has improved from lower impedance toward 50 Ohm. Thecorresponding Smith Chart also shows that the original response curvemoved toward 50 Ohm at the center of the Chart. Even though the curveslightly changed due to the termination with the synthesized matchingnetwork LPF1, the overall input impedance did not appear to change much.

Further, FIG. 10D shows the response of the original modulator MZM withthe 50 Ohm load only, e.g., dBS21, without the synthesized matchingnetwork LPF1. The −3 dB frequency of dBS21 is a bit less than 10 GHz andthe −6 dB frequency is about 25 GHz. It is noted that in contrast to theoriginal response, FIG. 10E shows a significant improvement. That is,the response with the synthesized matching network LPF1 and 50 Ohm loadshown in FIG. 10E shows a significant improvement made from 10 GHz to 20GHz.

In the example, it is noted that the −3 dB frequency of the system withthe synthesized matching network LPF1 (e.g., dBS43) is located at around20 GHz. Further, FIG. 10F shows corresponding group delay (GD) responsesof the system without the synthesized matching network LPF1 (e.g.,dBS21) and with the synthesized matching network LPF1 (e.g., dBS43). Asshown in FIG. 10F, the group delay variations have significantlyimproved from 13 ps max to 7 ps max over the entire frequency range. Asa result, in the example, over the frequency range of 20 GHz, the groupdelay variation of the system may be reduced to less than half, comparedto the one without the synthesized matching network LPF1. This furthermeans that the linearity of the modulator MZM may be significantlyimproved, in an aspect of the present disclosure.

As a result, in various aspects of the present disclosure, by using thepresent technology, e.g., novel design methodology of an impedancematching network applicable to modulators, for example, Mach-ZehnderModulators (MZMs), a wideband matching network may be implemented tofurther improve system performance including the frequency response.Thus, in an aspect of the present disclosure, the inductance of a LPFbased matching network (that is designed through the impedance matchingnetwork) may be designed to absorb the inductance of bonding wires ofthe modulator including the MZM as part of the LPF, thereby mitigating(or even eliminating) degradation of a frequency response of themodulator MZM, e.g., degradation due to the adverse effects of thebonding wires. Further, by employing a LPF type matching network betweenthe output of the modulator MZM and the termination, in addition to thesignificant improvement in the bandwidth of the modulator MZM, asignificant reduction in variation of a group delay over the modulationbandwidth of the modulator MZM may be achieved.

In another aspect of the present disclosure, different methodologies maybe implemented to absorb the inductance of bonding wires at an output ofa modulator. By way of example, there may be various ways to absorb theinductance of bonding wires at the output of MZM LPF. One approach mayprovide, in an aspect of the present disclosure, that a combinedinductance of L1 and L4 in FIG. 9 may be implemented as the inductanceof the bonding wire. Then, the capacitance C1 and inductance L3 may beimplemented in the submount of the modulator MZM. In another aspect ofthe present disclosure, the second approach may provide that a dummyinductance (L1+L4) and a dummy capacitance C1 may be implemented on themodulator MZM output port and the inductance L3 may be used to absorbthe inductance of the bonding wire.

Further, in the event that the implementation of dummy components on themodulator MZM output port is difficult, a different type of LPF having1^(st) order or 2^(nd) order filter LPF matching network may also besynthesized to make the methodology simple. Furthermore, it is notedthat the inductance of the bonding wire may be realized by adjusting thelength and shape of the bonding wire. If the inductance of thesynthesized inductor is too small, it may be difficult to implement.Therefore, some attention may be paid to make sure that the synthesizedinductance has a proper implementable value as a bonding wire.

In another aspect of the present disclosure, to additionally decrease oreliminate the adverse effects of the bonding wires of the modulator MZM,in addition to the output matching network (or a termination network) asdescribed herein, an input matching network may also be implemented,alone or in combination with the output matching network. By way ofexample, FIG. 11 illustrates one example design that includes an inputmatching network 1101 (e.g., LPF2) as well as an output matching network1103 (e.g., the termination network LPF1). The Butterworth LPF may bedisposed between the input matching network LPF2 and the output matchingnetwork LPF1 for the modulator MZM. In the example, modulator inputimpedance may be about 30 Ohm and it may be well maintained even with atermination network attached at an output of the modulator 1105, whichmeans that the input impedance of the Butterworth LFP may also be keptaround 50 Ohm.

As such, in the example shown in FIG. 11 , two possible ways may beavailable to synthesize the input matching network 1101 LPF2 for inputmatching: Case 1 and Case 2. In particular, for Case 1 (a case of a 50Ohm/30 Ohm output), the input matching network 1101 LPF2 may be matchedto 30 Ohm impedance. In this case, the input impedance of the LPF2 maybe 50 Ohm and the output impedance of the LPF2 may be 30 Ohm. Further,in Case 1, the inductance of 0.1231 nH of the Butterworth LPF may beabsorbed as part of the LPF2.

As for Case 2 (a case of a 50 Ohm/50 Ohm Output), the input matchingnetwork 1101 LPF2 may be matched to 50 Ohm impedance. In this case, theinput and output impedance may be 50 Ohm. Thus, the inductance of 0.1231nH of the Butterworth LPF may need to be kept as-is as part of theButterworth LPF to keep the input impedance of 50 Ohm.

Example implementation design methodology for input matching for Case 1and Case 2 may be illustrated as follows.

Case 1: Design of Input Matching LPF (50 Ohm/30 Ohm Output). FIG. 12illustrates an example of a schematic diagram for the design of theinput matching network LPF in an aspect of the present disclosure. It isnoted that the input matching network LPF2 1101 as shown in FIG. 11needs to have a very wide bandwidth in order to pass all input signaltoward the modulator (e.g., MZM) 1105. As such, in one implementationand in an aspect of the present disclosure, the bandwidth of the inputmatching network LPF2 1101 may be chosen to have about five (5) timesthe bandwidth of the modulator 1105 (e.g., MZM). In the example, a3^(rd) order Butterworth LPF may be selected for the input matchingnetwork LPF2 and each component thereof may be synthesized. Whensynthesizing the input matching network LPF2 1101 as shown in FIG. 11 ,since the input inductance L11 (i.e., 0.1231 nH) is part of theButterworth LPF that is synthesized for the modulator 1105, an outputinductance of the input matching network LPF2 1101 may absorb the inputinductance L11 (e.g., 0.1231 nH) of the Butterworth LPF as part of theinput matching network LPF2 1101. As such, the output inductance that issynthesized needs to be at least the same as the input inductance valueL11 (e.g., 0.1231 nH) or be greater than the input inductance value L11(e.g., 0.123 nH).

FIGS. 13A-13D show simulation results of Case 1 input matching networkof FIG. 12 . In FIGS. 13A-13D, dashed lines (e.g., dBS33) representresults of a design without an input matching network LPF2 (e.g., LPF21101), and solid lines (e.g., dBS55) represent results a design with aninput matching network LPF2 (e.g., LPF2 1101). As shown in FIG. 13A, inthe example, the input reflection is improved around up to 17 GHz andslightly degraded at above 17 GHz. The corresponding Smith Chart shownin FIG. 13B shows that input impedance matching has improved at a lowerfrequency side and that at higher frequency side the input impedance ismoving to a higher impedance area in the Smith Chart. Further, FIG. 13Cshows the frequency responses of two systems—one without the inputmatching network (i.e., dBS43) and the other with the input matchingnetwork (e.g., dBS65), which are almost identical to each other up to 17GHz. However, at above 17 GHz, in the example of FIG. 13C, the frequencyresponse with the input matching network LPF2 shows a slightdegradation. Further, as for the group delay, as shown in FIG. 13D thetwo group delay curves (one for with the input matching network LPF2 andthe other for without the input matching network LPF2) are adjusted tohave the same values at low frequency (e.g., below 5 GHz) and showvariations at high frequency (e.g., greater than 5 GHz). As such, it isnoted that with the input matching network LPF2, the group delay isshown to have about 1 ps degradation at high frequency end. Nonetheless,over most of the frequency band, the group delay is shown to haveimproved by 1 ps up to around 22 GHz.

Case 2: Design of Input Matching LPF (50 Ohm/50 Ohm Output). FIG. 14illustrates an example of a schematic diagram for the design of inputmatching network LPF in an aspect of the present disclosure. Asmentioned above, the input matching network LPF2 1101 may need to have avery wide bandwidth in order to pass all input signal toward themodulator 1105. Thus, in an aspect of the present disclosure, in asimilar manner to those with respect to FIG. 12 , the bandwidth of theinput matching network LPF2 1101 may be chosen to have about five (5)times that of the modulator 1105. In the example, a 3^(rd) orderButterworth LPF may be chosen for the input matching network LPF2 andeach component of the LPF2 may be synthesized as illustrated in FIG. 14.

Simulation results of Case 2 may be shown as in FIGS. 15A-15D. In thesimulation results, the dashed lines (e.g., dBS33) represent a designwithout the input matching network LPF2 1101, and the solid lines (e.g.,dBS77) represent a design with the input matching network LPF2 1101. Asshown in FIG. 15A, around up to 15 GHz, the input reflection hasimproved but slightly degraded at above 15 GHz. The corresponding SmithChart shown in FIG. 15B illustrates that the input matching has improvedat lower frequency but at higher frequency the input impedance is movingto a higher impedance area in the Smith Chart. It is also noted that theinput matching of Case 1 is slightly better than the one of Case 2.Further, as shown in FIG. 15C, the frequency responses of bothdesigns—one with the input matching network LPF2 (e.g., dBS43) and theother without the input matching network LPF2 (e.g., dBS87) are shown tobe almost identical to each other up to 15 GHz. At above 15 GHz,however, the frequency responses slightly degrade. Further, FIG. 15Dshows that the group delay curves for the group delay performance areillustrated to have substantially the same value at low frequency, butat high frequency, some variations. In particular, with the inputmatching network LPF2, the group delay is shown to have degraded lessthan 1 ps at high frequency end. However, over most of the frequencyband, the group delay is shown to have improved by 1 ps up to around 22GHz. As such, in comparison of Case 1 and Case 2, it may be noted thatthe amount of group delay variation of Case 2 is smaller than that ofCase 1, which may be due to the degradation at high frequency end.

In an aspect of the present disclosure, for the input matching networkLPF2, as shown in FIG. 11 , a Butterworth type filter may be chosenarbitrarily with caution to absorb the inductance of 0.1231 nH (e.g.,L11 or L17) of the Buttterworth LPF, which is an input part of themodulator 1105. Overall, as in Case 1 and Case 2, both designs show verysimilar input reflections, frequency responses and group delayperformance. As such, in an aspect of the present disclosure, insynthesizing the input matching network LPF2 1101, either approach(e.g., Case 1 or Case 2) may be adopted.

Also, it is noted that the inductance L15 in FIG. 12 , the inductanceL19 and L20 in FIG. 14 correspond to a bonding wire between themodulator and the submount. Further, it is noted that parts such asinductors, capacitors, etc. can be implemented in the submount. Thus, byimplementing the input matching network LPF2, in an aspect of thepresent disclosure, the effect of the inductance of bonding wires (e.g.,the inductance L15 in FIG. 12 , the inductance L19 and 20 in FIG. 14 )may be absorbed in the input matching network LPF2 1101. As a result, awideband modulator may be implemented without having any adverse effectdue to the bonding wires of the modulator.

As mentioned above, the present technology described herein provides anovel design methodology for mitigating or reducing the effect ofbonding wires. As discussed, often during packaging of MZM devices onsubmounts, bonding wires are inevitable components and it is noted thatthe inductance of the bonding wires degrade the frequency response ofthe MZM devices at high frequencies. This degradation especially becomesserious when the bandwidth of the MZM devices becomes very wide, forexample, a bandwidth of more than 30 GHz. The present technologydisclosed herein thus provides new, novel design methodologies tomitigate or reduce the effect of the bonding wires, thereby furtherimproving the system characteristics (e.g., frequency response and/orgroup delay response) of the MZM devices at high frequencies.

In accordance with various aspects of the present technology, low passfilter (LPF) type circuit topology may be adopted for a widebandmatching at an input and/or an output of a MZM on the submount to reduceor eliminate the adverse effects of the bonding wires between the MZMand the submount. That is, for a very wideband matching at the inputand/or output of the MZM, low pass filter type matching networks may bedesigned and implemented to mitigate the adverse effects of theinductance of the bonding wires, thereby improving the frequency andgroup delay responses of the MZM at high frequencies, e.g., a bandwidthof more than 20 GHz.

Further, the adverse effects of bonding wires (e.g., extra inductance)at the input and/or the output of the MZM may be absorbed as part of thelow pass filter type matching network. Furthermore, a low pass filtertype matching network placed at the output of the MZM may providespecial benefits. By way of example, an output matching network mayimprove frequency responses (S21) of the MZM significantly. In oneexample disclosed herein, the improved frequency response of the MZMresulted in a bandwidth gain of more than 2 times the original −3 dBbandwidth of the MZM. Also, the group delay variation may significantlyimprove. That is, the group delay variation in a system may besignificantly reduced, i.e., by at least 50% of the original MZMmodulator. Especially, smaller group delay may provide key benefits forthe modulation of higher order modulation (HOM) signals such as PAM4 orPSKs. The reduction in the group delay is achieved by the matchingnetwork by packaging and electrical characteristics of the MZM itself.However, it is noted that the present technology does not improve thedistortion caused by electrical to optical (EO) conversion process.

As such, a systematic synthesis methodology to absorb the effect ofbonding wires may be introduced to avoid performance degradations due tothe bonding wires between the MZM and the submount. In one embodiment,an L-C-L type equivalent circuit of MZM may be used to form aButterworth low pass filter (LPF) to convert the MZM impedance to 50Ohm. Further, as disclosed herein, two design approaches for the inputmatching may be used: one for matching to 50 Ohm of Butterworth MZM LPF,and the other for matching directly to the characteristic impedance ofMZM. Additionally, the matching network to 50 Ohm of Butterworth MZM LPFmay be disclosed for the output matching.

Further, instead of making an equivalent circuit of MZM and synthesizinga LPF network matching to 50 Ohm, input and/or output matching networksmay be synthesized directly to the MZM characteristics impedance. Also,an arbitrary L-C matching network other than LPF may be used when therequired frequency response of MZM is not very wide. As long as asynthesized matching network provides serial inductances at the inputand the output with proper inductance values, those inductances may beimplemented using bonding wires which may be absorbed as part of thematching networks.

In various aspects of the present disclosure, the present disclosureprovides one or more systematic approaches of designing input and/oroutput matching networks for MZM to reduce or eliminate the effect ofthe inductance of bonding wires, as well as to significantly improve thefrequency response and/or group delay characteristics, thereby improvingthe bandwidth or the frequency response of the MZM at high frequencies.

As used in the present disclosure, except explicitly noted otherwise,the term “comprise” and variations of the term, such as “comprising,”“comprises,” and “comprised” are not intended to exclude otheradditives, components, integers or steps.

The terms “first,” “second,” and so forth used herein may be used todescribe various components, but the components are not limited by theabove terms. The above terms are used only to discriminate one componentfrom other components, without departing from the scope of the presentdisclosure. Also, the term “and/or” used herein includes a combinationof a plurality of associated items or any item of the plurality ofassociated items. Further, it is noted that when it is described that anelement is “coupled” or “connected” to another element, the element maybe directly coupled or directly connected to the other element, or theelement may be coupled or connected to the other element through a thirdelement. A singular form may include a plural form if there is noclearly opposite meaning in the context. In the present disclosure, theterm “include” or “have” used herein indicates that a feature, anoperation, a component, a step, a number, a part or any combinationthereof described herein is present. Further, the term “include” or“have” does not exclude a possibility of presence or addition of one ormore other features, operations, components, steps, numbers, parts orcombinations. Furthermore, the article “a” used herein is intended toinclude one or more items. Moreover, no element, act, step, orinstructions used in the present disclosure should be construed ascritical or essential to the present disclosure unless explicitlydescribed as such in the present disclosure.

Although the present technology has been illustrated with specificexamples described herein for purposes of describing exampleembodiments, it is appreciated by one skilled in the relevant art that awide variety of alternate and/or equivalent implementations may besubstituted for the specific examples shown and described withoutdeparting from the scope of the present disclosure. As such, the presentdisclosure is intended to cover any adaptations or variations of theexamples and/or embodiments shown and described herein, withoutdeparting from the spirit and the technical scope of the presentdisclosure.

What is claimed as:
 1. A method of absorbing inductance comprising:using a low pass filter (LPF) type matching network configured to absorbinductance of bonding wires at a high frequency between an externaloptical modulator and a submount in a packaging of the external opticalmodulator.
 2. The method of claim 1, wherein the LPF type matchingnetwork comprises a first matching network coupled to an input of theexternal optical modulator and a second matching network coupled to anoutput of the external optical modulator.
 3. The method of claim 1,wherein the external optical modulator comprises a Mach ZehnderModulator (MZM).
 4. The method of claim 1, wherein the high frequencycomprises a frequency greater than 10 GHz.
 5. The method of claim 1,wherein the LPF type matching network comprises a 3^(rd) orderButterworth filter.
 6. The method of claim 1, wherein the high frequencycomprises a frequency comprises a frequency greater than 10 GHz.
 7. Themethod of claim 1, wherein the LPF type matching network comprises afirst matching network coupled to the output port of the MZM and whereinthe first matching network includes a resistor, an inductor, and acapacitor and is configured to absorb inductance of the bonding wires.8. The method of claim 7, wherein the LPF type matching networkcomprises a second matching network coupled to the input port of the MZMand the second matching network includes an inductor, and a capacitorand is configured to absorb inductance of the bonding wires.
 9. Themethod of claim 1, wherein the LPF type matching network comprises afirst matching network coupled to the output port of the MZM and asecond matching network coupled to the input port of the MZM, andwherein the first matching network and the second matching networkrespectively comprises an inductor and a capacitor and is configured toabsorb inductance of the bonding wires.